Automatic calibration of field-oriented elevator motor drive parameters using standstill motor measurements

ABSTRACT

An elevator controller 7 is provided with logic 48 which automatically calculates a motor time constant (τ R ), a torque constant (K T  *), and a magnetizing current (Id) for a field-oriented motor/drive system by calculating a transient inductance Lσ, injecting a variable input frequency sinewave into a q-axis reference current I qREF  and varying the input frequency while calculating an imaginary part of a rotor impedance Imag(Z R ) to obtain a frequency (F PEAK ) at which the maximum value of Imag(Z R ) occurs, then calculating τ R  from F PEAK . K T  *, Id, and a total motor voltage V M  are calculated and Id is varied until a ratio of rated motor voltage (Vph --  RATED) to V M  is within a predetermined tolerance of one. The procedure is performed with the rotor locked.

CROSS REFERENCES TO RELATED APPLICATIONS

Co-pending U.S. patent applications, Ser. Nos. 08/996,234, 08/996,263,08/996,262, 08/996,266, 08/996,264, filed contemporaneously herewith,contain subject matter related to that disclosed herein.

CROSS REFERENCES TO RELATED APPLICATIONS

Co-pending U.S. patent applications, Ser. Nos. 08/996,234, 08/996,263,08/996,262, 08/996,266, 08/996,264, filed contemporaneously herewith,contain subject matter related to that disclosed herein.

TECHNICAL FIELD

This invention relates to calibrating a field-oriented motor drive andmore particularly to automatic calibration of field-oriented (orvector-controlled) drive parameters for an elevator motor drive.

BACKGROUND OF THE INVENTION

It is known in the art of field-oriented (or vector-controlled) motordrives and motor speed controls that such drives and controls requireknowledge of the motor parameters such as the rotor time constant(τ_(R)), torque constant (K_(T) *), and rated magnetizing currentI_(dRATED).

One technique used to determine these motor parameters is to analyze themotor in an engineering laboratory using a dynamometer and expensivetest equipment and performing time consuming, highly technical tasks bya skilled technician or engineer. However, in modernization or retrofitapplications, where a new drive replaces an older drive in an existingelevator system, it is not convenient or cost effective to remove themotor from the elevator system for evaluation of the motor.

Also, it is desirable to determine the motor parameters based onmeasurements made on the motor at standstill. While techniques exist todetermine the motor parameters while the motor is running (at no-loadand under-load), it is not always practical to run such tests in anelevator application. In particular, a no-load test is not practicalbecause this would require unroping the elevator or disconnecting themotor from the gearbox. Further, performing a test under load, i.e.,with the motor connected to an elevator, is not practical because it isnecessary to have approximately correct motor parameters to start theelevator moving to obtain measurements of the motor running under-load.Also, it is desirable that the technique for determining such motorparameters be wholly contained within the drive control itself, so thatfield commissioning of retrofit or modernization drives may be performedby installers and service personnel without requiring specificmotor/drive tuning skills.

DISCLOSURE OF THE INVENTION

Objects of the invention include provision of automatic, on-site,calibration of motor parameters for field-oriented drives and/orcontrols for elevators, which does not require removal or uncoupling ofthe motor from the elevator system, and which uses only standstillmeasurements of the motor.

According to the present invention, a method for calculating at leastone parameter of an elevator motor operated by field-oriented control,using standstill measurements of the motor, the motor having a motorimpedance Z_(M), a rotor impedance Z_(R), and a transient inductance Lσ,includes: a) providing a sinusoidal torque current reference signal at ahigh frequency (F_(HIGH)) high enough such that the motor impedance(Z_(M)) is dominated by the transient inductance (Lσ); b) measuring afeedback torque current (Iq) and a feedback torque voltage (Vq); c)calculating the transient inductance (Lσ) at the high frequency(F_(HIGH)), by the equation: Lσ=Imag (Z_(M))@F_(HIGH) /(2πF_(HIGH)),where Z_(M) =Vq/Iq; d) providing a sinusoidal torque current referencesignal having a variable input frequency; e) measuring the feedbacktorque current (Iq) and the feedback torque voltage (Vq); f) calculatingan imaginary part of the rotor impedance Imag(Z_(R)) as follows:Imag(Z_(R))=Imag(Z_(M))-ωLσ, where ω is the input frequency, and Z_(M)=Vq/Iq; g) varying the input frequency and performing steps (d)-(f) toobtain the frequency (F_(PEAK)) at which the maximum value ofImag(Z_(R)) occurs; and h) calculating a rotor time constant (τ_(R))based on F_(PEAK).

According further to the present invention, τ_(R) is calculated asfollows: τ_(R) =1/(2πF_(PEAK)). According still further to the presentinvention, an additional step of calculating a magnetizing inductance Lφis performed as follows: Lφ=2Imag(Z_(R))/ω@ω=1/τ_(R).

According still further to the present invention, the followingadditional steps are performed: i) calculating a motor torque constant(K_(T) *) using the equation: K_(T) *=(3/2)(P/2)LφId*, where: P=numberof motor poles, Id*=Vph₋₋ RATED/(ωR₋₋ RATED×Lφ), and Vph₋₋ RATED =ratedmotor voltage, ωR₋₋ RATED=rated motor speed; j) calculating a motorvoltage (Vm*) using the equation: Vm*=(Vd*² +Vq*²)^(1/2), where:Vd*=ω_(E) LσIq*, Vq*=ω_(E) LsId*, Iq*=T₋₋ RATED/K_(T) *, T₋₋ RATED isthe rated motor torque, ω_(E) *=ω_(R).sbsb.--_(RATED) +ω_(S) * , andω_(S) *=(1/τ_(R)) (Iq*/Id*); k) calculating a ratio of Vph₋₋ RATED toVm*; and 1) varying Id* and performing steps (h)-(k) until the ratio iswithin a predetermined tolerance of 1.

The present invention represents a significant improvement over theprior art by providing automatic calibration of a field-oriented (orvector-controlled) induction motor controller for an elevator systembased on standstill measurements of the induction motor. The inventionprovides motor parameters such as the rotor time constant (τ_(R)),torque constant (K_(T) *), and rated magnetizing current I_(dRATED),without disconnecting the motor from the elevator system or from thegearbox. The invention also computes transient inductance Lσ,magnetizing inductance Lφ, stator resistance Rs, and rated torquecurrent I_(qRATED), as needed by a given application. Further, theinvention does not require a specially trained engineer with specialtest equipment to tune the motor/drive system. Thus, the inventiongreatly reduces cost associated with tuning the motor drive when newmotors drives are retrofit into job sites. Accordingly, automaticcalibration of motor parameters at the field site saves both time andmoney. As a result, the present invention makes it more attractive forbuilding owners to upgrade their elevator systems to modern controls,which are currently economically impractical due to the high cost ofdetermining parameters of older motors found in modernization job sites.

The foregoing and other objects, features and advantages of the presentinvention will become more apparent in light of the following detaileddescription of the exemplary embodiments thereof, as illustrated in theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic block diagram of a motor controller withauto-calibration logic, in accordance with the present invention.

FIG. 2 is a schematic block diagram of a current regulator/motor drive,in accordance with the present invention.

FIG. 3 is a schematic drawing of an equivalent circuit model of aninduction motor controlled by field orientation, in accordance with thepresent invention.

FIG. 4 is a simplified schematic diagram of the equivalent circuit ofFIG. 3, in accordance with the present invention.

FIG. 5 is a logic flow diagram of the auto-calibration logic of FIG. 1,in accordance with the present invention.

FIG. 6 is a graph of the imaginary part of the rotor impedance and ofthe motor impedance versus frequency, in accordance with the presentinvention.

FIG. 7 is a logic flow diagram of a portion of the auto-calibrationlogic of FIG. 5, in accordance with the present invention.

BEST MODE FOR CARRYING OUT THE INVENTION

Referring to FIG. 1, a portion of an elevator motor controller 7 shownto the left of the line 9, includes a field-oriented (or vector-based)motor control that has two control loops each corresponding to adifferent control axis, a d-axis relating to motor magnetization, and aq-axis relating to torque. The d-axis loop has a d-axis currentreference input signal I_(dREF) provided on a line 14. I_(dREF) is setto a predetermined constant value so as to provide appropriate magneticflux in the motor based on motor magnetization curves, e.g., I_(dRATED)or I_(NO-LOAD), discussed more hereinafter. The I_(dREF) signal is fedto a field-oriented current regulator/motor drive circuit 20, describedmore hereinafter with FIG. 2.

The q-axis current loop has a first q-axis current reference inputsignal I_(qREF1) on a line 15 is fed to one input of a switch 19.I_(qREF1) is provided by other logic (not shown), such as speed loopcompensation logic (not shown), which closes a motor speed control loop,such as that described in Copending U.S. patent application, Ser. No.(Otis Docket No. OT-3054), which provides the q-axis current referencesignal to the controller when it is not in auto-calibration.

The other input to the switch 19 is a second q-axis current referenceinput signal I_(qREF2) on a line 17. The output of the switch 19 is theq-axis current loop reference signal I_(qREF) on a line 18, which is setequal to I_(qREF1) or I_(qREF2) based on the state of the MODE1 signalprovided to the switch 19 on the line 13. The I_(qREF) signal is fed tothe field-oriented current regulator/motor drive circuit 20, describedmore hereinafter with FIG. 2.

Two examples of three phase AC induction motors used with the presentinvention are, Model LUGA-225LB-04A, by Loher, having a rated power of45 KW, rated voltage of 355 volts, rated speed of 1480, and ratedfrequency of 50 Hz, in a geared configuration; and Model 156MST, byTatung (of Taiwan), having a rated power of 40 KW, rated voltage of 500volts, rated speed of 251, and rated frequency of 16.7 Hz, in a gearlessconfiguration. Other motors having other rated parameters may be used ifdesired.

The circuit 20 provides three phase voltage signals Vx,Vy,Vz on lines 22to a motor 24, e.g., a three phase induction motor. The motor 24 isconnected by a mechanical linkage 26, e.g, a shaft and/or a gearbox, toa sheave 28. A rope or cable 30 is wrapped around the sheave 28 and hasone end connected to an elevator car 32 and the other end connected to acounterweight 34. The weight of the counterweight is typically equal tothe weight of an empty car plus 40-50% of the max load in the car.

Other elevator system configurations, and with or without acounterweight, with or without a gearbox, may be used if desired toconvert the output torque of the motor 24 to movement of the elevatorcab 32, such as dual lift (where two elevator cars are connected to asingle rope, the cars move in opposite directions and each car providesa counterweight for the other car), drum machine (where the rope iswrapped around a drum driven by a motor), etc.

A brake 37, e.g., an electromagnetic actuated disk brake, is disposed onthe shaft 26, and is driven by an electrical brake command signal BRKCMDon a line 38 from the circuit 20. The brake 37, when activated or"dropped", clamps onto the shaft 26 and prevents the motor shaft 26 fromturning, i.e., locks the rotor, and thus prevents the sheave 28 frommoving.

Referring to FIG. 2, it is known in the art of field-oriented motorcontrol that such control uses current (Id,Iq) and voltage (Vd,Vq)parameters corresponding to the d and q axes, respectively. Using fieldorientation, the motor magnetic field (or flux) will be controlled by Idand the motor torque will be controlled by Iq, as is known. Inparticular, the field-oriented current regulator/motor drive 20 of FIG.1 comprises two current control loops, one for the d-axis current Id andone for q-axis current Iq. The Id loop receives the I_(dREF) signal onthe line 14 to a positive input to a summer 102. A measured or feedbackd-axis current signal Id on a line 104 is fed to a negative input to thesummer 102. The output of the summer 102 is an error signal I_(dERR) ona line 106 which is fed to control compensation logic 108, such asproportional-plus-integral (P-I) current loop control. Other currentloop control compensation may be used if desired. The logic 108 providesa d-axis voltage command signal V_(dCMD) on a line 110.

For the q-axis, the Iq loop receives the I_(qREF) signal on the line 18to a positive input to a summer 114. A measured or feedback q-axiscurrent signal Iq on a line 116 is fed to a negative input to the summer114. The output of the summer 114 is an error signal I_(qERR) on a line118 which is fed to control compensation logic 120, e.g.,proportional-plus-integral (P-I) logic similar to the logic 108. Theoutput of the logic 120 is a q-axis voltage command signal V_(qCMD) on aline 122. Other control compensation, e.g., proportional, lead-lag,etc., may be used for the logics 108,120. The form of compensation usedis not critical to the present invention.

The voltage commands V_(dCMD) and V_(qCMD) are fed to knownfield-oriented to three-phase conversion logic 124 which converts thed-axis and q-axis voltage commands to three phase voltage commandsV_(XCMD), V_(YCMD), V_(ZCMD) on lines 126. The phase voltage commandsV_(XCMD), V_(YCMD), V_(ZCMD) are fed to a known three phase drivecircuit (or inverter) 128 which provides three phase voltagesV_(X),V_(Y),V_(Z) on lines 130, 132, 134, respectively (collectively,the lines 22), to drive the motor 24.

Within the drive circuit 128 (details not shown), each of the voltagecommands V_(XCMD), V_(YCMD), V_(ZCMD) on lines 126 are converted topercent duty cycle commands indicative of the corresponding inputvoltage level. The percent duty cycle is converted into apulse-width-modulated drive signal which drives power transistors toprovide the pulse-width-modulated, variable frequency, three phasevoltages V_(X),V_(Y),V_(Z) on lines 130, 132, 134, respectively. Theconversions within the drive 128 are performed using electroniccomponents and/or software well known in the art of motor drivecircuits. Any other type of drive circuit that receives input voltagecommands and provides output phase voltages may be used, and the phasevoltages need not be pulse-width modulated.

Phase currents I_(X), I_(Y), I_(Z) associated with the voltagesV_(X),V_(Y),V_(Z), respectively, are measured by known current sensors136, 138, 140, e.g., closed-loop Hall-effect current sensors (such asLEMS), respectively, and are provided on lines 132, 134, 136,respectively. The phase currents I_(X), I_(Y), I_(Z) are fed to knownthree phase to field oriented conversion logic 142, which provides aknown conversion from phase currents to d-axis and q-axis currents onthe lines 104, 116 which are fed to the summers 102,114, respectively.

The converters 124,150 provide known conversions between vector (d and qaxis) parameters and per-phase parameters, such as that described in D.Novotny, et al, "Vector Control and Dynamics of AC Drives", OxfordUniversity Press, 1996, Ch 5, pp 203-251. The converters 124,15 maylikely implement such conversions in software using a microprocessor orthe like.

It is known in the art of field oriented drives that the value of therotor time constant τ_(R) of the motor being controlled is required toperform the conversion to and from the field oriented d and q axes. Inparticular, τ_(R) is used to establish the correct slip frequency ω_(S)to achieve field orientation. The value of the rotor time constant τ_(R)is provided to the two converters 124, 150 on a line 144.

The motor drive logic 111 also includes a brake drive circuit 145 whichreceives an input signal BRK on a line 146 and provides a BRKCMD signalon the line 38.

Referring to FIG. 1, the present invention comprises auto-calibrationlogic 48 which automatically computes the motor parameters τ_(R), K_(T)*, I_(dRATED), and provides τ_(R) on the line 144 to the circuit 20,provides I_(dRATED) as I_(dREF) on the line 14 to the circuit 20, andprovides K_(T) * on a line 160 to speed loop compensation logic (notshown) such as that described in Copending U.S. Patent Application(OT-3054), filed contemporaneously herewith. The logic 48 also computesother motor parameters such as transient inductance Lσ, magnetizinginductance Lφ, stator resistance Rs (or R1) and rated torque currentI_(qRATED). The logic 48 receives Vq and Iq from the circuit 20. Thelogic 48 also provides the current reference signal I_(qREF2) to theswitch 19, and provides I_(dREF) to the circuit 20 on the line 14.

The logic 48 comprises known electronic components, which may include amicroprocessor, interface circuitry, memory, software, and/or firmware,capable of performing the functions described herein.

The logic 48 also provides the MODE1 signal on the line 13 to the switch19. The MODE1 flag causes the current reference signal I_(qREF2) fromthe calibration logic 48 to be fed to the logic 20. The logic 48 alsoprovides a break request signal BRK on the line 146 to the circuit 20.

The calculation logic 48 also communicates with a service tool 80 over aserial link 82. The service tool 80 includes a display 84 and a keypad(or keyboard) 86 for entering data into the service tool 80 and over thelink 82 to the controller 7. In particular, the logic 48 receives astart command over the link 82 from the service tool 80, which controlswhen auto-calibration is started. The logic 66 also provides a DONEsignal and a FAULT signal to the service tool 80 over the link 82. TheDONE signal indicates when auto-calibration has completed without faultsand the FAULT signal indicates when a fault has been detected duringauto-calibration.

Referring to FIG. 3, a known equivalent circuit 90 of an induction motoris similar to that described in "Vector Control and Dynamics of ACDrives", Novotny and Lipo, Oxford 1996, Chapter 5. FIG. 3 is a per-phaseequivalent circuit for AC steady state operation where the current I1and voltage V1 are phasor quantities. The circuit 90 comprises aresistor Rs in series with an equivalent "transient" inductor Lσ inseries with a rotor impedance Z_(R) which comprises a "magnetizing"inductance Lφ in parallel with an equivalent resistance R₂ /S. Where:

Rs (or R₁)=stator winding resistance

Ls=stator winding inductance

Lr=rotor winding inductance

Lm=mutual inductance

Rr=the rotor winding resistance

Lσ=Ls-Lm² /Lr=transient inductance

Lφ=Lm² /Lr=magnetizing inductance

ω_(E) =electrical frequency of the input current I₁

ω_(R) =motor output rotational speed in radians per second referred toan electrical reference frame

S=Slip=(ω_(E) -ω_(R))/ω_(E)

ω_(S) =slip frequency=ω_(E) -ω_(R) =(1/τ_(R))(Iq/Id)

where τ_(R) =rotor time constant Iq=q-axis (or torque) current, andId=d-axis (or magnetizing) current

R₂ =(Lm² /Lr²)*Rr

Also, the rotor time constant τ_(R) and motor torque constant K_(T) *are related to the parameters of the circuit 90 as follows:

    τ.sub.R =Lr/Rr=Lφ/R.sub.2

    K.sub.T *=(3/2)(P/2)LφId=torque/current

where P=number of poles.

Referring to FIG. 4, the circuit 92 is an equivalent to the circuit 90of FIG. 3 with the rotor impedance Z_(R) transformed into an equivalentseries circuit impedance having a real part Real(Z_(R)) and an imaginarypart Imag(Z_(R)) equal to ωLx. The equivalent circuit 92 with thetransformation of Z_(R) is useful for determining the rotor timeconstant τ_(R) (discussed more hereinafter).

Referring now to FIG. 5, a top level flow chart for the logic 48 beginsat a step 200 which determines whether a start command has been receivedfrom the service tool 80. If it has not, the logic exits. If a startcommand has been received, a step 202 requests and receives motorparameters from the service tool 80 over the link 82 (FIG. 1), which areentered by service personnel. The motor parameters received are: therated motor shaft power in watts (PWR₋₋ RATED); the rated motor speed inrpm (RPM₋₋ RATED); the rated rms line-to-line voltage in volts (VLL₋₋RATED); the rated frequency in hertz (HZ₋₋ RATED); and number of poles(POLES), all of which may be obtained from the motor nameplate data.

Then, a step 203 sets MODE1=1, BRK=1 to cause the brake 37 (FIG. 1) tolock the rotor, and I_(dREF2) =0 amps. For each of the tests describedherein, the rotor remains locked (rotor speed ω_(R) =0) and I_(dREF2) =0amps. When ω_(R) =0 and I_(dREF2) =0, the slip S=1, and the motorcurrent I₁ is equal to the q-axis current Iq and the motor voltage V₁ isequal to the q-axis voltage Vq. When Iq=0, the motor is operated insingle phase operation, in accordance with the circuits of FIGS. 3,4.

Next, a step 204 measures the transient inductance Lσ, by providing asinusoidal current signal into the q-axis of the reference currentI_(qREF2) on the line 17 (FIG. 1) at a frequency F_(HIGH) high enoughsuch that the motor impedance will be dominated by the transientinductance Lσ, e.g., 31.25 Hertz. Other frequencies may be used ifdesired, e.g., greater than 30 Hz. The sinewave input signal isgenerated digitally by a signal processor, such as a digital signalprocessor, e.g., a Motorola DSP 56002 processor, with an update (orsample) rate of 5 KHz. Other hardware and/or software techniques orupdate rates may be used to generate the sinusoidal input signals.

The step 204 reads the q-axis feedback current Iq and the q-axis outputvoltage Vq (equal to the motor current I₁ and motor voltage V₁,respectively, as discussed hereinbefore). Next, the step 204 uses theaforementioned digital signal processor to perform a Discrete FourierTransform (DFT) of Iq and Vq to determine the first harmonic Fouriercoefficients. The fundamental or first harmonic component of a measuredsignal from a DFT is A sin(ωt)+B cos(ωt), where ω=2πf is the inputfrequency (in rad/sec). The first harmonic is used to calculate theimpedance primarily so that non-linearities in the system do not distortthe calculation.

To compute a DFT, as is known, standard sine and cosine waves of unitamplitude at the test frequency are generated within the logic 48. Themeasured signal (Iq,Vq) is multiplied by the standard sinewave and theproduct is integrated over one period of the excitation to yield theFourier series coefficient A of the signal. Multiplying the signal bythe standard cosine and integrating yields the B coefficient. We havefound that integrating over 15 periods of the input signal it issufficient to filter out any transients in the system response. Othernumbers of periods may be used if desired. Also, for any DFT discussedherein, other types of Fourier transforms may be used if desired, e.g.,a Fast Fourier Transform (FFT), etc., provided the first harmonic of thedesired signal is obtained. Further, instead of a Fourier transform, anyother filtering or spectrum analysis technique may be used fordetermining the first harmonic of the desired signals.

Then, the step 204 computes the motor impedance Z_(M) by calculating theratio of voltage to current (V₁ /I₁ =Vq/Iq) using the first harmoniccomponents of voltage and current computed above. The step 204 thencomputes the real and imaginary parts of Z_(M) from the Fouriercoefficients. The imaginary part of the motor impedance Z_(M) atF_(HIGH) Hertz is dominated by the transient term ωLσ. Thus, thetransient inductance Lσ is the transient reactance (or imaginary part ofZ_(M)) with the input frequency equal to F_(HIGH) Hz, divided by thefrequency ω in radians/sec (2πF_(HIGH)), or:

    Lσ=Imag(Z.sub.M)@F.sub.HIGH Hz/(2πF.sub.HIGH)

Next, an optional step 206 measures the total resistance of the circuitimpedance (R_(TOT) =R_(s) +R₂), i.e., the sum of the stator and rotorresistances, as the real part of the motor impedance Z_(M) determined instep 204. Thus:

    R.sub.TOT =Real(Zm)@F.sub.HIGH

In particular, at the relatively high frequency F_(HIGH) used in step204, the inductance Lφ in the circuit 90 is large and the real part ofZ_(M) will be equal to R_(TOT). The value of R_(TOT) is saved for lateruse to calculate Rs (see step 212).

Next, a step 208 measures the rotor time constant τ_(R) as follows. Thestep 208 produces a progression of low frequency sinusoidal input q-axisreference currents I_(qREF2) from 0.1 to 8.0 Hertz in increments definedby a search algorithm, discussed hereinafter. The sinewave input signalis generated digitally as discussed hereinbefore with step 204. At eachfrequency, the motor current Iq and voltage Vq (equal to the motorcurrent I₁ and voltage V₁, respectively, as discussed hereinbefore) aremeasured and a DFT of the current I₁ and the motor voltage V₁ signalsare computed separately. The fundamental or first harmonic Fouriercoefficients are obtained as discussed hereinbefore with step 204.

Then, the step 208 computes the motor impedance Z_(M) at each frequencyby calculating the ratio of voltage to current (V₁ /I₁). The step 208then calculates the real and imaginary parts of the Z_(M) from theFourier coefficients. Then, the step 208 calculates the imaginary partof the rotor impedance Imag(Z_(R))=ωLx by subtracting the transientreactance (ωLσ) from the imaginary part of the motor impedance Z_(M),where Lσ was previously calculated in step 204 and ω is the inputfrequency, as follows:

    Imag(Z.sub.R)=ωLx=Imag(Z.sub.M)-ωLσ

Referring to FIG. 6, a curve 250 shows the imaginary part of the motorimpedance Imag(Z_(M))=ω(Lσ+Lx) and a curve 252 shows the imaginary partof the rotor impedance Imag(Z_(R))=ωLx. The frequency ω (radians/second)at which the maximum 254 of the curve 252 occurs is the inverse of therotor time constant, i.e., ω=1τ_(R). A known search algorithm, e.g., a"golden section line search" algorithm, varies the input frequency anddetermines the frequency Fpeak at which the maximum value of ωLx occurs.The type of search algorithm used is not critical to the presentinvention, and any search algorithm that varies an input parameter anddetermines the maximum value of an output parameter may be used. Therotor time constant τ_(R) is then calculated as follows:

    τ.sub.R =1/ωpeak=1/(2πFpeak)

Next a step 210 calculates the magnetizing inductance Lφ. In particular,at the frequency of rotor time constant (ω=1/τ_(R)) which is also thebreak frequency of the motor transfer function, the real and imaginaryparts of the rotor impedance Z_(R) are equal to each other, i.e.,ω(L_(x) =R_(x). Also, at this same frequency, it can be shown (below)that the ωL_(x) is also equal to 1/2ωLφ (the magnetizing reactance). Inparticular, the rotor impedance Z_(R) is equal to jωLφ in parallel withR₂, as shown below:

    Z.sub.R =jωLφR.sub.2 /(R.sub.2 +jωLφ)

Multiplying the numerator and denominator by the complex conjugate ofthe denominator (R₂ -jωLφ), gives

    Z.sub.R =ω.sup.2 Lφ.sup.2 R.sub.2 /(R.sub.2.sup.2 +ω.sup.2 Lφ.sup.2)+jωLφR.sub.2.sup.2 / (R.sub.2.sup.2 +ω.sup.2 Lφ.sup.2)                                             Eq. 1

Which has the form of a series combination of impedances, or a real andimaginary part as indicated below:

    Z.sub.R =Rx+jωLx

    Z.sub.R =Real+j Imaginary

At the peak 254 of the curve 252 of Imag(Z_(R)), the real and imaginaryparts are equal, which gives:

    ω.sup.2 Lφ.sup.2 R.sub.2 /(R.sub.2.sup.2 +ω.sup.2 Lφ.sup.2)=ωLφR.sub.2.sup.2 / (R.sub.2.sup.2 +ω.sup.2 Lφ.sup.2)                                             Eq.2

Simplifying Eq. 2, gives:

    ωLφ=R.sub.2

Substituting R₂ =ωLφ into the Imaginary part of Z_(R), and setting equalto ωLx, gives:

    Imag(Z.sub.R)=(ωLφ)(ω.sup.2 Lφ.sup.2)/(ω.sup.2 Lφ.sup.2 +ω.sup.2 Lφ.sup.2) =ωLx      Eq. 3

Simplifying Eq. 3, gives:

    ωLx=ωLφ/2                                  Eq. 4

Thus, the magnetizing inductance Lφ is calculated as follows:

    Lφ=2Imag(Z.sub.R)/ω@ω=1/τ.sub.R

Next, an optional step 212 calculates the stator resistance R_(S) byfirst calculating the value of R₂. It can be shown (below) that the realpart of the rotor impedance Real(Z_(R)) at ω=1/τ_(R) is equal to R₂ /2.In particular, the real part of Eq. 1 is:

    Real(Z.sub.R)=Rx=Lφ.sup.2 R.sub.2 /(R.sub.2.sup.2 +ω.sup.2 Lφ.sup.2)

Substituting R2=ωLφ, and simplifying, gives:

    Rx=R.sub.2 /2

Thus,

    R.sub.2 =2Real(Z.sub.R)@ω=1/τ.sub.R

Alternatively, R₂ may be calculated using the equation:

    R.sub.2 =Lφ/τ.sub.R

where Lφ and τ_(R) were previously calculated in steps 204,208,respectively. In either case, the stator resistance R_(S) is thendetermined by subtracting R₂ from the total resistance (R_(TOT) =R_(S)+R₂) calculated in step 206. Thus,

    R.sub.S =R.sub.TOT -R.sub.2

If the value for R_(S) for the motor is already known, e.g., from thedata sheet, it may be provided to the control over the link 82 and thenR_(S) may also be range checked in the step 212 to ensure it is within apredetermined percentage of the expected value. If R_(S) is not withinthe desired range, the step 212 sets a fault flag FAULT=1.Alternatively, the value of R_(S) may be calculated and provided to theservice tool to help service personnel determine the type of motorinstalled in the system.

Next, a step 214 uses Lφ, τ_(R), and the input parameters PWR₋₋ RATED,RPM₋₋ RATED, VLL₋₋ RATED, HZ₋₋ RATED, and POLES, obtained in the step202, to simulate motor parameters and to iterate and calculate the ratedmagnetizing current I_(dRATED), and the torque constant K_(T) *, asshown in FIG. 7.

Referring to FIG. 7, the simulated motor parameters are indicated by anasterisk (*) to avoid confusion with actual measured motor parametersdiscussed hereinbefore. In particular, a step 300 calculates the ratedrotational speed of the motor referred to the electrical reference frameω_(R).sbsb.-- RATED. Next, a step 302 converts rated line-to-linevoltage (VLL₋₋ RATED) to rated line-to-neutral voltage (or per-phasevoltage) Vph₋₋ RATED. Next, a step 303 calculates the rated torque T₋₋RATED based on rated power and rated RPM. Then, a step 304 calculatesthe stator inductance Ls as the sum of the transient inductance Lσ andthe magnetizing Lφ. Next, a step 306 calculates an initial value forsimulated d-axis current Id* based on a first order approximation of Idusing rated voltage and speed. Next, a step 308 sets a variable COUNTequal to zero.

Next, a series of steps 310-322 calculates K_(T) * and a simulated motorvoltage V_(M) * using various simulated motor parameters based on thevalue of Lφ calculated in step 210 (FIG. 5), the parameters calculatedin steps 300-308 above, and using known relationships for afield-oriented motor controller, some of which are discussedhereinbefore. In particular, a step 310 calculates the torque constantK_(T) * based on Lφ calculated in step 210 (FIG. 5) and the currentvalue of the magnetizing current Id*. Next, a step 312 calculates thetorque current Iq*. Next, a step 314 calculates a simulated slipfrequency ω_(S) * which is used in a next step 316 to calculate asimulated electrical current frequency ω_(E) * which is equal to therotational frequency (or speed) of the motor (emulated as the ratedspeed) ω_(R).sbsb.-- RATED plus the slip frequency ω_(S) *. Next, a step318 calculates a simulated q-axis output voltage Vq* based on themagnetizing current Id* and a step 320 calculates a simulated d-axisoutput voltage Vd* based on the torque current Iq*. Then, a step 322calculates a simulated vector sum total motor voltage Vm* equal to thesquare root of the sum of the squares of the d-axis and q-axis outputvoltages Vd*,Vq*, respectively.

Next, a step 324 calculates a Ratio parameter equal to the ratio of therated phase voltage Vph₋₋ RATED to the simulated per-phase motor voltageVm*. The logic iterates until the Ratio goes to 1 within the desiredtolerance, e.g., 0.001. When the ratio equals 1 the value of Id* willproduce the rated voltage at the rated RPM and rated torque.

Next, a step 326 calculates a next value for Id* equal to the value ofthe Ratio times the current value of Id*. Next, a step 328 checkswhether Ratio is within a predetermined tolerance of 1, e.g., 0.001. Ifit is not within the desired tolerance, a step 330 checks whether COUNTis greater than or equal to ten (i.e., whether the loop has iterated atleast ten times). If the loop has iterated at least ten times, a FAULTflag is set equal to 1 at a step 332 and output to the service tool 80over the link 82 (FIG. 1) and the logic is exited. If it has iteratedless than ten times, a step 334 increments COUNT by one and the logic214 goes the step 310 to iterate again.

If Ratio is within the desired tolerance in step 328, the logic isdeemed to have converged and at convergence the values of Id* and Iq*are equal to the rated d-axis current I_(dRATED) and the rated q-axiscurrent I_(qRATED), respectively. Accordingly, a step 340 sets thed-axis current reference I_(dREF) equal to Id* which is equal toI_(dRATED) and a step 344 sets I_(qRATED) equal to Iq*. Then the logic214 exits and returns to the logic 48 of FIG. 5.

Referring to FIG. 5, next, a step 216 determines whether an error hasbeen detected in any of the above steps 202-214 (i.e., if FAULT1=1). Ifa fault has been detected, a step 218 sets FAULT=1 which is sent to theservice tool 80 (FIG. 1) over the serial link 82 and a step 220 setsMODE1=0, BRK=0, and the logic exits. If a fault has not occurred, a step222 sets the DONE flag equal to 1 which is transmitted via the seriallink 82 to the service tool 80. Next, some or all of the motorparameters τ_(R), K_(T) *, I_(dRATED), Lσ, Lφ, Rs, and I_(qRATED) aretransmitted via the serial link 82 to the service tool 80 in a step 224.The service tool 80 displays the parameters for use by the servicepersonnel. Next, the step 220 sets MODE1=0, BRK=0, and the logic 48exits.

Although the invention has been described and illustrated with respectto exemplary embodiments thereof, it should be understood by thoseskilled in the art that the foregoing, and various other changes,omissions and additions may be made without departing from the spiritand scope of the present invention.

What is claimed is:
 1. A method for calculating at least one parameterof an elevator motor operated by field-oriented control, usingstandstill measurements of the motor, the motor having a motor impedanceZ_(M), a rotor impedance Z_(R), and a transient inductance Lσ,comprising the steps of:a) providing a sinusoidal torque currentreference signal at a high frequency (F_(HIGH)) high enough such thatthe motor impedance (Z_(M)) is dominated by the transient inductance(Lσ); b) measuring a feedback torque current (Iq) and a feedback torquevoltage (Vq); c) calculating said transient inductance (Lσ) at said highfrequency (F_(HIGH)), by the equation:

    Lσ=Imag(Z.sub.M)@F.sub.HIGH /(2πF.sub.HIGH)

where Z_(M) =Vq/Iq; d) providing a sinusoidal torque current referencesignal having a variable input frequency; e) measuring the feedbacktorque current (Iq) and the feedback torque voltage (Vq); f) calculatingan imaginary part of the rotor impedance Imag(Z_(R)) as follows:

    Imag(Z.sub.R)=Imag(Z.sub.M)-ωLσ

where ω is said input frequency, and Z_(M) =Vq/Iq; g) varying said inputfrequency and performing steps (d)-(f) to obtain the frequency(F_(PEAK)) at which the maximum value of Imag(Z_(R)) occurs; and h)calculating a rotor time constant (τ_(R)) based on F_(PEAK).
 2. Themethod of claim 1, wherein said τ_(R) is calculated as follows:

    τ.sub.R =1/(2πF.sub.PEAK).


3. The method of claim 1, wherein said step of calculating Zm comprisescomputing a first harmonic of Vq and a first harmonic of Iq.
 4. Themethod of claim 3, wherein said step of computing the first harmonic ofIq and Vq, comprises computing Fourier transforms of Iq and of Vq. 5.The method of claim 2, further comprising a step of calculating amagnetizing inductance Lφ as follows:

    Lφ=2Imag(Z.sub.R)/ω@ω=1/τ.sub.R.


6. The method of claim 5, further comprising the steps of:i) calculatinga motor torque constant (K_(T) *) using the equation:

    K.sub.T *=(3/2)(P/2)LφId*

where: P=number of motor poles;

    Id*=Vph.sub.-- RATED/(ωR.sub.-- RATED×Lφ);

and Vph₋₋ RATED=rated motor voltage; ωR₋₋ RATED=rated motor speed; j)calculating a motor voltage (Vm*) using the equation:

    Vm*=(Vd*.sup.2 +Vq*.sup.2).sup.1/2

where: Vd*=ωLσIq*; Vq*=ω_(E) LsId*; Iq*=T₋₋ RATED/K_(T) *; T₋₋ RATED isthe rated motor torque; ω_(E) =ω_(R--RATED) +ω_(S) *; and ω_(S)*=(1/τ_(R))(Iq*/Id*); k) calculating a ratio of Vph₋₋ RATED to Vm*; andl) varying Id* and performing steps (h)-(k) until said ratio is within apredetermined tolerance of
 1. 7. The method of claim 6, wherein saidstep of varying Id* comprises calculating the next value of Id* asId*×Ratio.
 8. The method of claim 5, further comprising the steps of:m)calculating a total resistance (RTOT) at said high frequency, by theequation:

    R.sub.TOT =Real(Zm)@F.sub.HIGH

n) calculating a stator resistance (Rs) as follows:

    R.sub.S =R.sub.TOT -R.sub.2

where R₂ =Lφ/τ_(R) ; and o) performing step (m) between the steps (b)and (d).
 9. The method of claim 1, wherein said steps (a)-(h) areperformed automatically upon receiving a command from a service tool.10. The method of claim 6, wherein said steps (a)-(1) are performedautomatically upon receiving a command from a service tool.